Isolated current monitoring circuit for measuring direct and high duty factor currents

ABSTRACT

A method and circuit for measuring direct and high duty factor current in a conductor (1) with minimal interference with the operation of a monitored circuit. Current flow in the conductor is magnetically sensed with a transformer T1 having a primary winding connected electrically in series with the conductor (1). The transformer T1 is driven into saturation during a first time interval and brought out of saturation during a second time interval. After the transformer T1 is brought out of saturation, an output signal is provided which is proportional to the flow of current in the conductor (1).

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to current monitoring circuits andsystems. More specifically, the present invention relates to analogcircuits and systems for monitoring current in electrically isolatedcircuits.

While the present invention is described herein with reference toillustrative embodiments for particular applications, it should beunderstood that the invention is not limited thereto. Those havingordinary skill in the art and access to the teachings provided hereinwill recognize additional modifications, applications, and embodimentswithin the scope thereof and additional fields in which the presentinvention would be of significant utility.

2. Description of the Related Art

For many applications, there is a need to determine flow of current in acircuit or conductor. The conventional approach involves the insertionof a series resistor in the conductor. However, the use of a resistor isproblematic, lossy and impractical in applications wherein anabove-ground voltage is present on the current carrying conductor. Moreimportantly, in some applications, the use of the resistor substantiallyinterferes with the intended operation of the circuit. Hence, for theseapplications, there is a need for an indirect (or isolated) method formonitoring the current flow in the conductor.

One conventional isolated current monitoring scheme involves use of Halleffect devices. A Hall effect device is a device, which is magneticallycoupled to the current carrying conductor. In order to accomplish therequired magnetic coupling, the Hall effect device must be placed inseries with the magnetic path which encircles the conductor. The Halleffect device has a typical thickness of 0.04 inch. Since the Halldevice has magnetic properties similar to those of air, an "air" gap iseffectively placed in the magnetic path. The low-permeability gapreduces the overall permeability of the magnetic circuit, to assure thatsaturation of the magnetic core is avoided. The inductor, thus formed,is effectively inserted in series with the current-carrying conductor ofthe current monitoring device. For some applications, this addedinductance is a circuit parasite capable of storing energy. Most Halleffect devices are prone to relatively large drifts in output whensubjected to temperature changes. Complex, and often troublesome,circuitry is frequently required to overcome this temperature driftproblem.

Another approach involves the use of a transformer to indirectly sensecurrent flow in the monitored circuit. The transformer provides a coil(typically a single-turn primary) to pick up energy in the magneticfield created by the flow of current through the monitored conductor orcircuit.

Transformers are often used for this purpose inasmuch as: 1) themagnetic coupling thereof provides electrical isolation from the circuitbeing monitored, 2) when large currents are to be measured, the use of asingle-turn primary and multi-turn secondary, reduces power loss fromthe circuit being monitored, and 3) proper choice of the transformerturns ratio provides improved signal-to-noise ratio for increasedaccuracy and trouble-free operation.

Despite these advantages, conventional transformer based currentmonitoring circuits cannot be utilized to measure current in DC (directcurrent) circuits or in those switched circuits that employ duty ratiossubstantially greater than 50%. This is due to the limitation thattransformers fail to function when the magnetic cores thereof aresaturated. Hence, conventional transformer based current monitoringcircuits provide for a transformer core "reset" during a required OFFportion of every cycle to ensure that the transformer does not "walk"into saturation.

Thus, there is a need in the art for a current monitoring circuit thatallows for the measurement of DC and high duty cycle currents withoutinterfering with the operation thereof.

SUMMARY OF THE INVENTION

The need in the art is addressed by the present invention which providesa method and circuit for measuring direct and high duty factor currentin a conductor. Current flow in the conductor is magnetically sensedwith a transformer having a primary winding connected electrically inseries with the conductor. The transformer is driven into saturationduring a first time interval and brought out of saturation during asecond time interval. After the transformer is brought out ofsaturation, an output signal is provided which is proportional to theflow of current in the conductor. Measurement of DC and high duty factorcurrents is effected with minimal interference with the operation of themonitored circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1(a) is a schematic diagram of a conventional transformer coupledcurrent monitor.

FIG. 1(b) illustrates a typical control signal applied to the switch Q1'of the conventional current monitoring circuit of FIG. 1(a) at node A.

FIG. 1(c) illustrates a typical output signal provided by theconventional current monitoring circuit of FIG. 1(a) at node B.

FIG. 2 (a) is a schematic diagram of a first illustrative embodiment ofthe current monitoring circuit constructed in accordance with theteachings of the present invention.

FIG. 2(b) is a diagram showing the waveform of the saturate commandsignal applied to inverting input of the operational amplifier of thecurrent monitoring circuit of FIG. 2(a).

FIG. 2(c) is a diagram showing the waveform of the sample command signalapplied to the sample and hold circuit of the current monitoring circuitof FIG. 2(a).

FIG. 3(a) is a first alternative embodiment of the current monitoringcircuit of the present invention.

FIG. 3(b) is a diagram showing the waveform of the saturate commandsignal applied to the inverting input of the operational amplifier ofthe current monitoring circuit of FIG. 3(a).

FIG. 3(c) shows a waveform that would be applied to the sample and holdcircuit of the current monitoring circuit of the present invention totime the operation thereof.

FIG. 4(a) is a second alternative embodiment of the current monitoringcircuit of the present invention.

FIG. 4(b) is a diagram showing the waveform of the saturation commandsignal applied to the secondary winding of the transformer of thecurrent monitoring circuit of FIG. 4(a).

FIG. 4(c) is a diagram of the output of the current monitoring circuitof FIG. 4(a).

FIG. 5(a) shows a third alternative embodiment of the current monitoringcircuit 10 of the present invention.

FIG. 5(b) shows the waveform that appears at the output of the pulsewidth modulator IC employed in FIG. 5(a).

FIG. 5(c) shows the saturation command signal that is generated by theone shot employed in FIG. 5(a).

FIG. 5(d) shows the analog of the current being monitored.

FIG. 6(a) shows a fourth alternative embodiment of the currentmonitoring circuit of the present invention.

FIG. 6(b) shows the waveform of the sample signal applied to the currentmonitoring circuit of FIG. 6(a).

FIG. 6(c) shows the waveform of the saturation signal applied to thecurrent monitoring circuit of FIG. 6(a).

FIG. 6(d) shows the waveform of the output of the operational amplifierof the current monitoring circuit of FIG. 6(a).

FIG. 6(e) shows the waveform of the magnetic flux of the transformer ofthe current monitoring circuit of FIG. 6(a).

FIG. 6(f) shows the waveform of the voltage on the secondary winding ofthe transformer of the current monitoring circuits of FIG. 6(a).

FIG. 6(g) shows the waveform of the current through the secondarywinding of the transformer of the current monitoring circuit of FIG.6(a).

FIG. 7(a) shows a fifth alternative embodiment of the current monitoringcircuit of the present invention.

FIGS. 7(b) and 7c) show the waveforms at the input and the output,respectively, of the waveform generator of the current monitoringcircuit of the present invention.

FIG. 8 shows a sixth alternative embodiment of the current monitoringcircuit of the present invention.

FIG. 9 is an illustrative embodiment of the Royer oscillator employed inthe current monitor of FIG. 8.

FIG. 10 shows the current monitoring circuit of the present inventionadapted to monitor the current applied by a missile control system to acontrol mechanism for a fin of a missile.

DESCRIPTION OF THE INVENTION

Illustrative embodiments and exemplary applications will now bedescribed with reference to the accompanying drawings to disclose theadvantageous teachings of the present invention.

FIG. 1(a) is a schematic diagram of a conventional transformer coupledcurrent monitor 10' capable of measuring current in a switched circuit(not shown) connected via a conductor 1. The primary winding of acurrent sense transformer T1' is connected in series with a power MOSFETswitch Q1'. The switch Q1' (usually part of the circuit being monitored)provides a chopping action that provides an off period in each cycleduring which the magnetic flux in the core of the transformer T1' isallowed to reset to near zero. During the off period of the switch, theminute magnetizing current that has accumulated in the current sensetransformer T1' during the on period is allowed to "dump" into zenerdiode VR1' via diode CR2 . The magnetic core of the current sensetransformer is thereby prevented from walking into saturation. WhenMOSFET Q1' conducts, a current is induced in the secondary winding oftransformer T1' that is in direct proportion to that in the primary.This current flows from the secondary through diode CR1', resistor R1'and returns to the secondary via ground. This produces a voltage acrossR1' that is directly related to the current in the primary. Diodes CR1'and CR2' thus steer the secondary current to R1' and VR1' respectivelyduring the appropriate portions of the cycle.

FIG. 1(b) illustrates a control signal applied to the switch Q1' at nodeA. FIG. 1(c) illustrates a typical output signal provided by theconventional current monitoring circuit of FIG. 1(a) at node B.

Although useful for some applications, this technique is not capable ofmeasuring direct current since the switch duty factor can not approach100% without causing the transformer core to saturate. This shortcomingis addressed by the present invention which provides a choppingtechnique that enables the circuit to monitor direct current. Thetechnique of the present invention involves deliberate saturation of thecore during a portion of the cycle in order to ensure that the core willnot be in saturation at the time when a data sample is to be captured.

FIG. 2(a) is a schematic diagram of a first illustrative embodiment ofthe current monitoring circuit 10 constructed in accordance with theteachings of the present invention. In the first illustrativeembodiment, the circuit invention 10 includes a transformer T1. Thetransformer core is typically a tape wound core made of very highpermeability magnetic material, of a type that has a very squarehysteresis loop to provide rapid saturation and de-saturation at theends of the hysteresis cycle. Examples of such magnetic materials aresquare Permaloy-80 manufactured by Magnetics Inc. of Butler,Pennsylvania. Also, certain rare earth materials could be used such asMetglass, manufactured by Allied Chemical Corporation. ("Metglass" is aregistered trademark of Allied Chemical Corporation.) The tape thicknessis typically 1 mil. (0.001 inch) or less. The transformer T1 has a oneturn primary winding connected in series with the conductor 1 from amonitored circuit (not shown). One end of the secondary winding of thetransformer T1 is connected to a circuit for driving the transformerinto saturation and bringing the transformer out of saturation (asdiscussed more fully below), while the second end of the secondarywinding is connected to a potential well (or ground). The first end ofthe secondary winding of the transformer is connected to a node at whichone end of first and second resistors R1 and R2 are connected to thecathode of a first diode CR1. The second end of the second resistor R2is connected to an inverting input of an operational amplifier U1 andthe second end of the first resistor R1 is connected to the output ofthe operational amplifier. A third resistor is connected between thenoninverting input of the operational amplifier U1 and the anode of thefirst diode CR1. A fourth resistor is connected between the noninvertinginput of the operational amplifier U1 and a source of positive supplyvoltage. The collector of a first (bipolar PNP) transistor Q1 is alsoconnected to the inverting input of the operational amplifier U1. Aclock signal is applied to the base of the first transistor Q1 by theclock 12. The emitter of the first transistor Q1 is connected to asource of positive bias voltage. The output of the operational amplifierU1 is connected to a sample and hold circuit 14.

In operation, the clock 12 applies a saturate command to the base of thefirst transistor Q1. When the saturate command signal goes low, thetransistor Q1 conducts and places a high at the inverting input of theoperational amplifier U1. FIG. 2(b) is a diagram showing the waveform ofthe saturate command signal applied to the inverting input of theoperational amplifier of the current monitoring circuit of FIG. 2(a).The chopping waveform, typically having 50% duty cycle, is applied tothe secondary of the current monitoring transformer T1, and consists ofthe following two states:

1) During state 1, the voltage applied to the transformer secondarywinding is of the same polarity as the minute voltage impressed acrossthe primary by the current being monitored. By assuring that thevolt-second product of this portion of the cycle is greater than that ofstate 2, the magnetic core is forced into saturation. This establishes aknown baseline condition, which ensures that the core cannot be drivento saturation in the opposite sense before the data is to be sampled.The choice of a core material that saturates abruptly offers theadvantage that when the applied voltage is reversed, it also unsaturatesabruptly, i.e., the current sensing transformer T1 quickly returns to alinear operating state.

2) The turn-off of Q1 initiates state 2, during which the choppingvoltage applied to the secondary is reversed from that of state 1, i.e.,it bucks and overrides the voltage produced by the current flowing inthe primary. Due to the squareness in the hysteresis loop materialemployed, the core is quickly returned to an unsaturated state by thisreversed voltage. The transformer T1 then functions as a linear currenttransformer for the remainder of state 2. The secondary winding currentis converted to a voltage and sampled in synchronism with the choppinginput. Such a circuit typically consists of an operational amplifier anda sample and hold integrated circuit. At the time the data samples arecaptured by the sample and hold circuit, the secondary current is 1/Ntimes the primary current, where N is the ratio of secondary turns toprimary turns. The gain of the circuit is controlled by the turns ratioof the transformer and the choice of feedback resistor employed with theoperational amplifier.

Turn-on of Q1 causes the signal at the inverting input to exceed that atthe noninverting input (e.g., 1 volt), the output of the operationalamplifier U1 goes low. With the operational amplifier U1 thus held in anopen loop state, the current flowing in the secondary of the transformerT1 flows through the first diode CR1. This is a return to saturationperiod during which the voltage at the ungrounded end of the secondarywinding is approximately -0.7 volts as determined by the drop across thefirst diode CR1. With a transformer turns ratio of 500:1, R1 equal to100 ohms, R2 and R3 equal to 4.02 K ohms each, R4 equal to 56.2 K ohmsand bias and supply voltages of ±15 volts applied to the operationalamplifier U1 and 15 volts applied to the fourth resistor R4, the voltageacross the single turn primary of T1 is less than 2 millivolts. After ashort period, perhaps a few microseconds, the first transformer T1returns to its saturated state and the voltage drop thereacross drops tozero where it remains until the next state 2 command.

State 2 is initiated by turning Q1 off. This allows the voltage on thecathode of the first diode CR1 to be delivered to the operationalamplifier U1 via the second resistor R2 and permits the amplifier loopto close. The voltage on the output of the operational amplifier U1rises to a level that causes the current in the secondary of thetransformer T1 to be delivered via the first resistor R1 from the outputof the operational amplifier U1. With the illustrative values set forthabove, the voltage across the secondary of the transformer T1 will nowbe at 1 volt as it is forced to match the 1 volt bias potential appliedto the noninverting input of the operational amplifier U1. This voltagereversal on the secondary of the transformer T1 causes the very squarehysteresis core of the transformer T1 to unsaturate quickly (e.g., in amicrosecond). As soon as the core unsaturates, the voltage at the outputof the operational amplifier U1 is:

    V.sub.out =V.sub.B +R1*I/N                                 [1]

where I is the current being monitored in the conductor 1, N is thetransformer turns ratio, and V_(B) is the bias voltage applied to thenoninverting input to the operational amplifier. This current monitoroutput pulse is then synchronously captured by a sample and hold circuit14 or an A/D converter (not shown). FIG. 2(c) shows a waveform thatwould be applied to the sample and hold circuit 14 to time the operationthereof. FIGS. 2(b) and 2(c) are in temporal alignment such that it isapparent that the sample pulse occurs just before the re-saturate signalgoes high.

Thus, the four part cycle of the present invention is 1) re-saturate, 2)rest in a saturated state, 3) unsaturate/linearize, and 4) sample.

FIG. 3(a) is a first alternative embodiment of the current monitoringcircuit of the present invention. This embodiment is essentially thesame as that of FIG. 2(a) with the exception that the transistor Q1 isreplaced by a second diode CR2, a buffer amplifier is provided by athird transistor Q3 and a fifth resistor R5 is added between the baseand the emitter of the third transistor Q3. In addition, a capacitor isprovided between the bias supplies. The input signal is again of theform shown in FIG. 2(b) while the sample signal is as shown in FIG.2(c).

In operation, a logic level chopping command is applied to the anode ofdiode, CR2. FIG. 3(b) is a diagram showing the waveform of the saturatecommand signal applied to the inverting input of the operationalamplifier of the current monitoring circuit of FIG. 3(a). The choppingwaveform, typically having a 50% duty cycle, is applied to the secondaryof the current monitoring transformer T1 and consists of the following 2states:

1) During state 1, the voltage applied to the transformer secondarywinding is of the same polarity as the minute voltage impressed acrossthe primary by the current being monitored. By assuring that thevolt-second product of this portion of the cycle is greater than that ofstate 2, the magnetic core is forced into saturation. This establishes aknown baseline condition, which ensures that the core cannot be drivento saturation in the opposite sense before the data is to be sampled.The choice of a core material that saturates abruptly offers theadvantage that when the applied voltage is reversed, it also unsaturatesabruptly, i.e., the current sensing transformer T1 quickly returns to alinear operating state.

When the chopping command signal transfers to its low state theoperational amplifier unsaturates, i.e., it is permitted to return tolinear operation. The output of the operational amplifier and itscurrent buffer, transistor Q3, supply a small voltage, 1 Volt, to thesecondary of transformer, T1, via resistor, R1. This voltage is of apolarity that bucks the voltage applied to the primary by the current tobe measured. Due to the squareness in the hysteresis loop materialemployed, the core is quickly returned to an unsaturated state by thisreversed voltage. The transformer T1 then functions as a linear currenttransformer for the remainder of state 2. The secondary winding currentis converted to a voltage and sampled in synchronism with the choppinginput. Such a circuit typically consists of an operational amplifier anda sample and hold integrated circuit or D/A converter. At the time thedata samples are captured by the sample and hold circuit, the secondarycurrent is 1/N times the primary current, where N is the ratio ofsecondary turns to primary turns. The gain of the circuit is controlledby the turns ratio of the transformer and the choice of feedbackresistor employed with the operational amplifier.

When the high state of the chopping command is applied to the anode ofdiode, CR2, the signal at the inverting input exceeds that at thenoninverting input (e.g., 1 volt), which causes the output of theoperational amplifier U1 to go low. With the operational amplifier U1thus held in an open loop state, the current flowing in the secondary ofthe transformer T1 flows through the first diode CR1. This is a returnto saturation period during which the voltage at the ungrounded end ofthe secondary winding is approximately -0.7 volts as determined by thedrop across the first diode CR1. With a transformer turns ratio of500:1, R1 equal to 50.0 ohms, R1 equal to 50.0 ohms, R2 and R3 equal to4.02 K ohms each, R4 equal to 56.2 K ohms, and bias and supply voltagesof +and -15 volts applied to the operational amplifier U1 and 15 voltsapplied to resistor R4, the voltage across the single turn primary of T1should be less than 2 millivolts. After a short period, perhaps a fewmicroseconds, the first transformer T1 returns to its saturated stateand the voltage drop thereacross drops to zero where it remains untilthe next state 2 command.

State 2 is initiated when the steering signal applied to the anode ofdiode CR2 goes low. This allows the voltage on the cathode of diode CR1to be delivered to the operational amplifier U1 via resistor R2 andpermits the amplifier loop to close. The voltage on the output of theoperational amplifier U1 rises to a level that causes the current in thesecondary of the transformer T1 to be delivered via resistor R1 from theoutput of the operational amplifier U1 and current buffer Q3. With theillustrative values set forth above, the voltage across the secondary ofthe transformer T1 will now be at 1 volt as it is forced to match the 1volt bias potential applied to the noninverting input of the operationalamplifier U1. This voltage reversal on the secondary of the transformerT1 causes the very square hysteresis core of the transformer T1 tounsaturate quickly (e.g., in a microsecond). As soon as the coreunsaturates, the voltage at the output of the operational amplifier U1is:

    V.sub.out =V.sub.B +R1*I/N                                 [2]

where I is the current being monitored in the conductor 1, N is thetransformer turns ratio, and V_(B) is the bias voltage applied to thenoninverting input to the operational amplifier. This current monitoroutput pulse is then synchronously captured by a sample and hold circuit14 or an A/D converter (not shown).

FIG. 3(c) shows a waveform that would be applied to the sample and holdcircuit 14 to time the operation thereof. FIGS. 3(b) and 3(c) are intemporal alignment such that it is apparent that the sample pulse occursjust before the re-saturate signal goes high.

FIG. 4(a) is a second alternative embodiment of the current monitoringcircuit of the present invention. In the embodiment of FIG. 4(a), thenoninverting input of the operational amplifier U1 is connected to aground reference and the inverting input is connected to the one end ofthe secondary winding of the transformer T1. The second end of thesecondary winding of the transformer is connected to the collector of afirst (bipolar NPN) transistor Q1. The emitter of the first transistoris connected to a -0.7 volt supply. The base of the first transistor Q1is connected to the collector of a second transistor Q2. In theillustrative embodiment of FIG. 4(a), the second transistor Q2 is a PNPbipolar transistor. The base terminal of the second transistor isconnected to ground.

The anode of a first diode CR1 is connected to the collector of thefirst transistor Q1 while the cathode of same is connected to theinverting input of the operational amplifier U1. Thus, the first diodeis connected between the ends of the secondary winding of thetransformer T1. A first resistor is also connected at one end to theinverting input of operational amplifier U1. A second resistor isconnected at one end to the output of the operational amplifier U1. Theoutput of the operational amplifier U1 is connected to the base of athird bipolar transistor Q3. The third transistor Q3 is a bipolar NPNtransistor in the illustrative embodiment of FIG. 4(a). The collectorterminal of the third transistor Q3 is connected to a source of a 5 voltsupply. The emitter of the third transistor Q3 is connected to thesecond ends of the first and second resistors R1 and R2 and provides theoutput of the current monitoring circuit 10.

A chopping input saturation signal, such as that illustrated in FIG.4(b), is applied to the transformer T1 through a third resistor R3, theemitter collector junction of the second transistor Q2 and the firsttransistor Q1.

The operation of the embodiment of FIG. 4(a) is essentially the same asthat of the embodiment of FIG. 2(a). The transistor Q2 serves to providea threshold for the activation of the first transistor Q1. The thirdtransistor Q3 buffers the output of the operational amplifier U1 andthereby provides increased current drive for the output therefrom. Theoutput of this circuit is depicted in FIG. 4(c) for an input signal ofthe form shown in FIG. 4(b).

In this circuit the operational amplifier U1 and its current buffertransistor Q3 function only as a current to voltage converter, with thechopping action being performed by first and second transistors Q1 andQ2. This eliminates the voltage pedestal from the output, so that theoutput is referenced to ground instead of 1 volt. In this thirdembodiment, the operational amplifier U1 remains linear throughout theentire cycle, i.e., a virtual ground is maintained on the cathode offirst diode CR1 throughout states 1 and 2. When the chopping signalsupplied to third resistor R3 goes low the second and first transistorsQ2 and Q1 cease conducting to initiate state 1. Current I flowingthrough the single-turn primary of transformer T1 induces a positivevoltage on the anode of CR1, causing CR1 to conduct. This current pathis from the secondary of T1 through CR1 and return to the secondary ofT1. The voltage developed across the secondary is limited by CR1 toapproximately 0.7 volts and the voltage across the primary is thuslimited to 0.7/N =1.4 millivolts. This voltage causes the transformer toreturn to saturation. When the transformer saturates the voltage acrossit drops to near zero where it remains for the balance of state 1.

State 2 is initiated when the clock/steering signal applied to thirdresistor R3 goes high. This causes current to flow through thirdresistor R3, second transistor Q2, and the emitter-base junction of thefirst transistor Q1, causing Q1 to conduct. The conduction of Q1 appliesapproximately -0.5 volts to the anode of first diode CR1 and thesecondary winding of transformer T1. This voltage is of a polarity thatbucks the voltage induced into T1 by the current I through the primary,and is opposite the polarity applied to the transformer during state 1.This reversed voltage causes the square hysteresis loop core of thetransformer to quickly unsaturate, returning it to a linear operatingstate. In this linear operating state current flows from operationalamplifier U1 and its buffer transistor Q3 through first resistor R1, thesecondary of transformer T1, collector to emitter of the firsttransistor Q1 to the -0.7 volt supply. Since the current through R1 isidentically the same as the current through the transformer secondary,and since one end of R1 is at virtual ground, the voltage V_(Q3) at theemitter of Q3 is:

    V.sub.Q3 =I*R1*1/N                                         [3]

Where: R1 is the resistance of R1 in ohms and N is the transformer turnsratio. In the example of FIG. 4(a) the voltage output is 0.2*I or 200mv/amp. In most applications this output voltage would be sampled with asample and hold integrated circuit or D/A converter near the end state2. State 2 ends when the clock signal goes low, and the cycle repeats.

FIG. 5 (a) shows a third alternative embodiment of the currentmonitoring circuit 10 of the present invention. FIGS. 5(b), 5(c) and5(d) respectively illustrate the waveforms which appear at the output ofthe pulse width modulator U1, the three microsecond ONE SHOT U3, and thecollector of Q1 in their proper time relationships to each other. Thisinvention is utilized in a buck converter to provide current feedback tothe pulse width modulator, without the restriction that switch dutyfactor cannot approach 100%. A secondary advantage of this configurationis that the minute inductance introduced into the circuit by the currentmonitor is not in series with the MOSFET switch. This helps to reducethe voltage spiking at the drain of Q3 when it turns off. This circuitprovides near-elimination of duty cycle restriction to realize increasedefficiency and improved tolerance to low input line voltage conditions.An integrated circuit could be produced that would combine the functionsof U1 and U3 into advanced configuration that would remove allrestriction from duty cycle. The result would be a slight additionalimprovement in dynamic range.

In operation, the output signal from U1, the pulse width modulator ICcontrols MOSFET Q3 and ONE SHOT IC U3. When Q3 turns off U3 istriggered, causing it to activate RESET transistor Q2. When Q2 isconducting, current flows from the +12 Volt housekeeping power supplythrough the secondary of current sense transformer T1, resistor R2,transistor Q2, and returns to the +12 Volt power supply via ground. Thevoltage impressed across the secondary of T1 is of the same polarity asthat produced by the current to be monitored, which flows in thesingle-turn primary. This voltage causes T1 to saturate within the threemicrosecond period allocated by ONE SHOT U3. State 2 begins when Q2 isturned off by U1 at the end of the three microsecond reset (state 1)interval. During state 2 the current flowing in the single-turn primaryof T1 causes a corresponding current, I_(PRI) /N, to flow in thesecondary. This current path is from the +12 Volt power supply throughthe secondary of T1, diode CR1, transistor Q1, resistor R1, and returnto the power supply via ground. This current, flowing through resistorR1, produces an analog feedback which is in proportion to the current tobe monitored. This voltage is supplied to the RAMP input to U1 viaresistor R6.

The waveforms of FIGS. 5(a), 5(b), and 5(c) appear at the respectivepoints indicated in FIG. 5. Resistive divider R7/(R7+R6) adds slopecompensation to this current analog. Slope compensation is required incurrent programmed buck converters operating in continuous inductorconduction mode with greater than 50% duty factor. This circuit thusemploys the saturating transformer current monitoring principle of thepresent invention to monitor inductor current with no significantlimitation on duty cycle. A secondary advantage is that the freewheeling rectifier recovery current spike does not pass through thesaturable transformer T1, thus eliminating the requirement for spikesuppression circuitry.

The DC-DC converter used in this embodiment is shown only as a block,which would be a typical load for the buck regulator. Resistor R8 andcapacitor C1 are generic, as required to determine the operatingfrequency. Capacitors C3, C4, and C5 are typical high Q bypasscapacitors. In the illustrative embodiment, resistor R5 is 15 ohms,resistor R3 is 5 K ohms, transistor Q1 is a 2N2907, transistor Q2 is a2N2222, MOSFET Q3 is a type IRF350, diode CR1 is a 1N4150, diode CR4 isa type UES804, inductor L1 is 400 microhenries, 10 Amp., and the pulsewidth modulator, U1, is a type UC1842 integrated circuit.

FIG. 6(a) shows a fourth alternative embodiment of the currentmonitoring circuit of the present invention. FIG. 6(b) shows thewaveform of the sample signal applied to the current monitoring circuitof FIG. 6(a). FIG. 6(c) shows the waveform of the saturation signalapplied to the current monitoring circuit of FIG. 6(a). FIG. 6(d) showsthe waveform of the output of the operational amplifier of the currentmonitoring circuit of FIG. 6(a). FIG. 6(e) shows the waveform of themagnetic flux of the transformer of the current monitoring circuit ofFIG. 6(a). FIG. 6(f) shows the waveform of the voltage on the secondarywinding of the transformer of the current monitoring circuit of FIG.6(a). FIG. 6(g) shows the waveform of the current through the secondarywinding of the transformer of the current monitoring circuit of FIG.6(a).

In FIG. 6(a), direct current flows in monitored conductor 102 and thesingle-turn primary 104 of the transformer 103 producing a minutevoltage across the primary that is positive at the end marked with apolarity dot. A waveform generator 121 applies the chopping waveform ofFIG. 6(b) to a resistor 107. During state 1, this input to the resistor107 is in its high state. At the beginning of state 1, there are twosignificant current flow paths on the secondary side of the transformer.The first is from waveform generator 121 through resistor 107, diode109, resistor 110, operational amplifier 111, and return to ground viathe negative supply 112. The second current path is from the secondary106 of the transformer 103 through a diode 109 and return to secondary106. The voltage across the secondary 106 of the transformer 103produced by the above-described currents flowing through diode 109 is ofsuch a polarity that it causes the magnetic flux in core 105 to returnto saturation. See the magnetic flux waveform of FIG. 6(e). When thetransformer 103 saturates, it becomes an effective short circuit acrossthe diode 109. For the remainder of state 1, current flows from thewaveform generator 121, resistor 107, secondary 106 of the transformer103, resistor 110, operational amplifier 111, and returns to ground viathe negative supply 112.

At the beginning of state 2, chopping waveform from the waveformgenerator 121 switches to its low state causing the diode 108 to startconducting. Current then begins to flow from the positive supply 115through operational amplifier 111, resistor 110, secondary 106 of thetransformer 103, and resistor 107 and returns to ground via waveformgenerator 121. The voltage across the secondary 106 of the transformer103 is thus reversed and held at 0.7 volts, by the forward conductionpotential of diode 108. This voltage is of a polarity that opposes thevoltage induced by current I flowing in the primary. After a short time,less than a millisecond, transformer core 105 is restored to itsunsaturated condition, so that the transformer 103 functions as a linearcurrent transformer for the remainder of state 2.

During this latter portion of state 2, when the transformer 103 isfunctioning as a linear current transformer, the input 116 to the sampleand hold circuit 117 is 1 volt/amp times the current I. This isdetermined by the turns ratio of the transformer 103 and resistor 110.With the illustrative component values shown, each ampere flowing in theprimary 104 of the transformer 103 causes 1/500 ampere to flow in thesecondary 106. This current 2 ma/amp flows through the resistor 110dropping 1 volt/amp thereacross. Since one end of resistor 110 is atvirtual ground at node 129, the output of the operational amplifier 111is identically 1 volt/amp. During this latter portion of state 2, whenthe output of the operational amplifier 111 has settled at a voltageproportional to the current in conductor 102, waveform generator 121supplies a sample command to sample and hold circuit 117 to updateoutput 119. Capacitor 118 holds this voltage until the next updateevent.

The transformer 103 used in this embodiment was constructed as follows.The core is part #80512-1/2 D MA supplied by Magnetics Inc., of Butler,Pa. This standard catalog item is a miniature tape wound toroid, 0.350inch diameter and 0.105 inch thick. The tape employed in the manufactureof the core was 0.5 mil (0.0005 inch) thick Square Permalloy 80, whichis an 80% nickel-iron Ni-Fe alloy. The secondary is 500 turns of AWG 36.magnet wire. The single-turn primary consists of the current-carryingAWG-18 wire passing through the center of the toroid. Diodes 108 and 109are 1N4150. Resistors 107 and 110 are 500 ohm 1/4 watt. Operationalamplifier 111 is one-half a TLO-72. Capacitors 113 and 114 are 0.1microfarad, 100 volt, ceramic. The waveform generator 121 and the sampleand hold circuit 117 were shown simplified in FIG. 6(a) for the purposeof illustration.

FIG. 7a shows a fifth alternative embodiment similar to that of FIG.6(a) with the exception that the reset and linearize voltages areapplied to the secondary of the saturable transformer in a somewhatdifferent manner. In operation, state 1 is initiated when the steeringvoltage output from wave form generator 121, goes high. Current thenflows from the waveform generator through diode 130, and forces theinverting input to operational amplifier 111, to go positive, with theresult that the operational amplifier output saturates in a negativedirection. This establishes a current path from ground through diodes108 and 109, resistor 110, operational amplifier 111, and returns toground via the negative supply, not shown. Two other current paths flowthrough resistors 122 and 123, but they are not important duringstate 1. The current flowing through diode 109, impresses a voltageacross secondary 106, of saturable transformer 105. This 0.7 volts is ofthe same polarity as that produced by the current I, which flows throughthe single-turn primary. After a short time, a millisecond or so, thetransformer is returned to saturation. The current previously flowingthrough diode 109 is then diverted through the secondary, since atransformer in saturation is essentially a short circuit. Thetransformer remains in saturation throughout the remainder of state 1.

State 2 is initiated when the steering output from waveform generatorgoes low, causing diode 130 to stop conducting. Current now flows fromthe positive supply, not shown, through operational amplifier 111,resistor 110, the secondary 106, of transformer 105, resistor 122, thenegative supply, not shown, and returns to the positive supply viaground. A second current path flows from ground through diode 108,resistor 122, and returns to ground via the negative supply. Thisapplies -0.7 volts to the end of the secondary that is marked with apolarity dot, and the operational amplifier applies virtual ground tothe opposite end of the secondary. The result is a small voltage 0.7volts, across the secondary that is of a polarity that bucks the voltageimpressed across the single-turn primary by current I. After a shorttime, less than a millisecond, the transformer is returned from itssaturated state to its linear operating state, at which time it beginsto function as a linear current transformer. A current path is thencaused to flow from the positive supply via operational amplifier 111,resistor 110, secondary 106, resistor 122, the negative supply, andreturns to the positive supply via ground. Since one end of resistor 110is at virtual ground, the output of operational amplifier 111 is seen tobe at:

    V.sub.o =I*R1/N.                                           [4]

This voltage is sampled near the end of state 2, in response to theSAMPLE command FIG. 7(c). The sampling device can be either a sample andhold IC, shown, or an A/D converter, not shown. For the parameters shownthe output is 2 Volts per Ampere.

The components used in this embodiment are essentially the same as thoseused in the previously described embodiments. The diodes are 1N4150,transformer 105 is identical to the one described for FIG. 2(a).Resistors 110 and 122 are 1.0 K ohm, 1/4 watt and resistor 123 is 2.0 Kohm, 1/4 watt. Operational amplifier 111, is one-half a TLO-72. Thewaveform generator and the sample and hold integrated circuit aregeneric devices.

FIG. 8 shows a sixth alternative embodiment of the present invention.This tandem embodiment is capable of supplying an essentially continuousoutput that is proportional to the DC current to be monitored. In brief,the transformer 122 is part of a square wave generator 124 which couldconsist of a Royer oscillator as shown in FIG. 9, or a Jensenoscillator.

Although not part of the invention, the functioning of the Royeroscillator of FIG. 9 is described for clarity as follows: Transformer100 is constructed with a saturable core made of the same or similarsquare hysteresis loop material as that used in the current monitortransformers of the current invention. When 15 volt power is applied,current initially flows through R3, tertiary winding 102, resistors R1and R2, and base to emitter in both transistors Q1 and Q2. This startsto turn both transistors on, but one will always prevail over the othercausing the one to saturate and the other to turn off. Assume that Q2prevails over Q1 initially. The conduction of Q2 causes the non polaritydot end of winding 101 of saturable transformer 100 to be clamped to theground potential. Transformer action then causes the non polarity dotend of tertiary winding 104 to go negative which turns Q2 off and thepolarity dot end of tertiary winding 104 to go positive, reinforcing theturn-on of Q2. This conduction-sustaining current path is from thepolarity dot end of tertiary winding 104 through resistor R2, base toemitter of Q2, ground, CR3, and return to the center tap of tertiarywinding 104. Resistor R3 is used only to provide a "tickle" of startupbias current. After startup a potential of minus 0.7 volts is maintainedat the cathode of diode CR3. During this half of the cycle the base ofQ1 is held approximately 4 volts negative, which maintains it in a nonconducting state. This initially described state is maintained untilsufficient volt-second product has been applied to the transformer todrive its core into saturation. When core saturation occurs, thetransformer primary loses its ability to maintain a bias currentsustaining voltage on the tertiary winding at which time this windingbecomes essentially a short circuit. This applies a ground potential tothe cathode of CR2 which quickly removes the stored charge from theemitter-base junction of Q2, causing it to turn off abruptly. With bothtransistors turned off, the magnetizing current in the transformerbegins to flow out of the non polarity dot end of tertiary winding 104into the base of Q1 via resistor R1. This turns Q1 on which initiatesthe second half cycle. This state will be maintained until the magneticflux in the transformer core reaches saturation in the opposite sense,and the cycle repeats. The Royer oscillator thus continues to oscillateand produce symmetrical square waves on all its transformer windings asthe saturable core is repeatedly driven from saturation to saturation.For the herein-described application, the two secondaries are used toprovide steering signals of opposite phases to the two push-pull halvesof the tandem version of the invention. These square wave signals areisolated from ground, so that virtual ground can be maintainedcontinuously by the operational amplifier used in the invention.Transformer secondaries 118 and 148 produce complimentary square wavesignals to provide the chopping excitation required to alternatelyrefresh and linearize two identical current monitors, of the type hereindescribed. Since the two current monitors are driven with opposite phasechopping signals, in a push-pull mode, a continuous current monitoroutput is realized.

The detailed operation is as follows. First with respect to the currentmonitor 117, the saturable transformer 105 is driven into saturation,its refresh condition, by secondary 120 of transformer 122 when thepolarity dot 121 is positive. Current flows from this positive end ofthe secondary 120 through a current limiting resistor 107, diode 109,diode 118 and returns to secondary 120. This applies a voltage acrosssecondary 106 of transformer 105 that is of the same polarity as thevoltage produced in primary 104 by current 101 in conductor 102. Thisdrives saturable transformer 105 into saturation to establish abaseline/refresh condition. When saturable transformer 105 saturates, itbecomes an effective short, so that the current previously flowingthrough diode 109 is diverted through secondary 106. This condition ismaintained for the balance of the half cycle during which polarity dot121 is positive.

When the chopping square wave polarity across winding 120 of transformer122 reverses, current flows from this secondary through diode 119, diode108, current limiting resistor 107, and returns to secondary 120. Sincethe anode of diode 108 is at ground potential, and this diode isconducting the above described current, it is evident that the cathodeof this diode, and one end of the secondary 106 of the transformer 105,are at a potential of approximately minus 0.7 volts. The other end ofsecondary 106 is at virtual ground 129. This establishes a voltageacross secondary 106 equal 0.7 volts and of such a polarity that itopposes the excitation produced by the current flowing the primary 104,i.e., the voltage applied to the secondary bucks that in the primary.This forces transformer 105 out of saturation and causes it to operateas a linear current transformer. This second current path flows from the+15 volt supply 115 through the operational amplifier 111, resistor 110,secondary 106 of the transformer 105, current limiting resistor 107,secondary 120 of the transformer 122, diode 119 and returns to powersupply 115 via the circuit ground. Since the current flowing in thesecondary 106 is the primary current divided by the turns ratio, andsince this same current flows through resistor 110, the voltage acrossresistor 110 is seen by Ohm's law to be in proportion to the current I.One end of the resistor 110 is at virtual ground 129 so that the outputvoltage is identically the voltage across resistor 110 and isproportional to the current I. Transformer 105 is Wound on a core thatsaturates and unsaturates abruptly. The transition from saturation, thereset condition, to linear operation occurs very quickly, typically afew microseconds.

The operation of the chopped current monitor 147 is identical to that ofthe chopped current monitor 117 described above. The chopped currentmonitors are operated in a push-pull phase relationship so that, whenone is being refreshed, the other is delivering an output. Therespective roles of the chopped current monitors are reversed during thenext half cycle, so that a near continuous output is produced withslight perturbations, at the transitions of the chopping square wavesproduced by square wave generator 124. Capacitors 126 and 125 andresistor 123 smooth these imperfections to a degree that would besatisfactory for most applications.

In this embodiment of the invention, the square chopping voltagessupplied by secondaries 120 and 150 are 6 volts peak and RMS. A typicalsquare wave frequency is 2.0 KHz. Diodes 108, 109, 118, 119, 138, 139,147 and 148 are type 1N4150. Resistors 107 and 137, are 120 ohm, 1/4watt. Capacitors 113 and 114 are 0.1 microfarad. With these parameterschosen, the gain of the current 25 monitor output 116 divided by input101 is 1.0 volt per amp over the useful range of zero to ten amps. Arequirement for the design of the circuit is that the current throughresistors 107 and 137 be greater than the maximum current through theresistor 110.

Those skilled in the art will appreciate the utility of the presentinvention. For example, the present invention provides an improvedmissile control apparatus. That is, FIG. 10 shows the current monitoringcircuit of the present invention 10 adapted to monitor the currentapplied by a missile control system 13 to a control mechanism 15 for afin (not shown) of a missile 11 via the conductor 1.

Thus, the present invention has been described herein with reference toa particular embodiment for a particular application. The use of a veryhigh permeability core material operates to assure that the errorintroduced by the magnetizing current is negligible for all but the mostcritical applications. The use of a very square hysteresis loop corematerial operates to assure that the minute magnetizing current islinear during the critical portions of the cycle. Compensating the errorintroduced by the minute linear magnetizing current is thereforeaccomplished simply, by a small increase, e.g., less than 1%, in theresistance of the operational amplifier feedback resistor.

Those having ordinary skill in the art and access to the presentteachings will recognize additional modifications applications andembodiments within the scope thereof.

It is therefore intended by the appended claims to cover any and allsuch applications, modifications and embodiments within the scope of thepresent invention.

Accordingly, What is claimed is:
 1. An isolated current monitoringcircuit for measuring direct and high duty factor current in a conductorcomprising:A. transformer means for magnetically sensing current flow insaid conductor, said transformer means including a transformer having aprimary winding connected electrically in series with said conductor anda secondary winding; B. means for driving said transformer intosaturation during a first time interval including:a. an operationalamplifier with an inverting input connected to the first end of saidsecondary winding of said transformer, a noninverting input connected tomeans for providing a reference potential and an output terminalconnected via a first resistor to the first end of the secondary windingof said transformer; and b. means connected to said inverting input ofsaid operational amplifier for providing a saturation control signal; C.means for bringing said transformer out of saturation and operation saidtransformer as a current transformer during a second time interval,comprising:c. said noninverting input connected to means for providing areference potential and an output terminal of said operational amplifierconnected via a first resistor to the first terminal of the secondarywinding of said transformer; and d. a second resistor connected to saidinverting input of said operational amplifier at one end and the firstend of said secondary winding of said transformer at a second endthereof; D. said means for providing a reference potential comprises:e.a third resistor connected to said noninverting input of saidoperational amplifier at one end and a potential well at a second endthereof; and f. a fourth resistor connected to said inverting input ofsaid operational amplifier at one end and a source of potential at asecond end thereof; and E. means for providing an output signal, afterthe initiation of the second time interval, which is proportional to theflow of current in said conductor.
 2. The invention of claim 1 whereinsaid means for bringing said transformer out of saturation during asecond time interval further includes a diode having a cathode terminalconnected to the first end of said secondary winding of said transformerand an anode terminal connected to said potential well.
 3. The inventionof claim 2 wherein said diode is connected between said second ends ofsaid second and third resistors.
 4. The invention of claim 3 whereinsaid second end of said secondary winding is connected to said potentialwell.